Modern switching-mode converters for power supplies, such as those used in personal computers and other electronic equipments are plagued by many practical problems.
First, for safety and other reasons, an isolation transformer is required in the power processing stage of the converters, which then requires an isolation transformer in the feedback control circuit leading to major circuit complexity and cost.
Second, for cost and size reasons, multiple output converters are preferred in a majority of practical applications to supply multiple voltages, such as +5 V, +12 V, and -15 V in a single converter. However, only one of the output voltages can be fully regulated, while the others cannot.
In addition, the load currents cannot be allowed to drop below certain minimum values (typically 15% of the full load value) without drastically affecting unregulated multiple output voltages. A typical solution to this problem has been to waste 15% of excess power by placing a resistor in parallel with the output load in order to maintain multiple outputs within prescribed tolerances.
Yet another problem is to shape the input current to follow the ac power line current in the power processing input stage of the converters for unity-power factor operation to avoid pollution of utility power lines with harmonic currents. Typically unity-power factor operation is obtained by use of an additional front-end switching-mode regulator, which increases both the size and cost of the power processing stage and decreases the power processing efficiency.
The nature of these practical problems is discussed more fully below with reference to FIGS. 1 through 5 using conventional switching-mode converters as examples.
A typical prior-art isolated switching-mode converter shown in FIG. 1 consists of an isolated power stage 10 and an integrated circuit (IC) feedback regulator 12, where the necessary regulation is effected. However, since the only controllable device, a switching transistor Q1, is on the primary side of an isolation transformer T1, the feedback circuit must also be isolated by a transformer T2. The single feedback regulator 12 comprises a voltage comparator 12a which compares the output voltage to a load with a reference voltage VREF, and a voltage comparator 12b which compares the error voltage from the comparator 12a with a sawtooth wave voltage generated at, for example, 20 kHz clock rate to produce a pulse width modulated (PWM) control signal to the switch Q1.
single integrated circuit (IC) regulator chips are commercially available from a number of manufacturers which perform all of the functions of the feedback regulator 12. Later figures include only the functional description of such a regulator chip in nonisolated feedback control circuits of this type.
The ac input line voltage is rectified by a full-wave diode rectifier shown for simplicity in the drawing as a single diode D but is in practice implemented with a diode bridge circuit. The rectified voltage is filtered by a capacitor C. The isolated switching-mode converter 10, comprising switching diodes D1 and D2 and inductive filter elements L1 and C1, is a conventional buck converter.
The feedback regulator 12 controls only one quantity, the duty cycle of the switching transistor Q1 on the primary side of the isolation transformer T1 Thus, both input voltage variations and output load current changes are absorbed by controlling the duty ratio of the single input switching transistor Q1.
The complexity of the circuit needed to implement isolation in the feedback loop is obvious from FIG. The control circuit on the secondary side of the isolation transformer T1 requires power, which is obtained from either a separate isolated switching-mode converter (not shown) or through a third, low power, isolation transformer T3 and a conventional linear power supply regulator 14 operated from the ac input line. However, the controlling switching transistor Q1 on the primary side of the isolation transformer then requires an isolation transformer T2 in the feedback loop. Thus, the need for some protection features on either side of the main isolation transformer T1 increases the number of isolation transformers to three because the linear power supply 14 required to provide regulated VCC for the feedback circuit 12 requires an additional isolation transformer.
As shown in FIG. 2, in multiple output converters, the single feedback control regulator 12 permits the full regulation of only the one output voltage shown in FIG. 1. The other output voltage can, depending on the load current, undergo very large variations from +12 V nominal value to as low as +6 V, or as high as +18 V, which is clearly unacceptable. The typical prior-art solution to this problem is to preload both outputs with some minimum 15% load current by use of dummy resistors in parallel with the load shown in FIG. 2 as being a variable resistor.
If the load RL in the main converter varies, as illustrated in FIG. 2, the output current i1 to the load varies, as shown by the dotted line in the graph of FIG. 2a, but the average voltage output to the load is maintained through the PWM of the duty cycle D to the switching transistor Q1 as intended. However, that change in duty cycle of the switch will then also have an effect on the output current to the load of the second of the multiple output converter. If, on the other hand, the load of the other of the two multiple outputs varies as shown in FIG. 3, it will be virtually unregulated as shown in FIG. 3, i.e., the duty cycle D is virtually unchanged, as will be understood more fully from further discussion below. While that technique of providing both outputs with some minimum 15% load current will alleviate the voltage regulation problem, it continuously wastes 30% of power and thus defeats the original purpose of using higher efficiency switching-mode regulators over linear-mode regulators.
Full load current regulation on multiple output voltages could, of course, be obtained as shown in FIG. 4 by use of the cascaded switching postregulators 16 and 18. Efficiency is reduced significantly since the total input power is processed twice, once in the main multiple-output switching-mode regulator 10' and the second time through the postregulators 16 and The size, weight and cost of the power-supply package is thus increased significantly.
Several important performance observations could be made with reference to the isolated Cuk converter 20 as disclosed in U.S. Pat. No. 4,184,197 and illustrated in FIG. 5. In this, as in all switching-mode converters used so far, the existence of the single controllable switching transistor Q on the primary side and the controlled converter 20 necessitates at least one level of isolation in the feedback circuit 22 for control of the output dc voltage on the secondary side. In practice, the isolation barrier is crossed many more times, such as in providing VCC for the feedback circuit. Note that the single active switch Q on the primary side performs a dual function, as in other switching dc-to-dc converters shown in FIGS. 1 and 2. Its pulse width is modulated to regulate against both the changes of input voltage and changes of output load current. Note also that of the two disturbances, input voltage variation has a more significant first order effect on duty ratio D, while load current changes cause only a small second order correction in the duty ratio.
For example, from the ideal voltage conversion gain (D/(1-D) without parasitic resistances, a 4 to 1 input voltage change (15 V to 60 V) may cause duty ratio change from 0.2 to 0.5 or 250% change. When the parasitic resistances are included in the voltage conversion ratio, the load current change by a factor of 10 (from 9% to 90% of load) typically may cause only 10% or less change of nominal duty ratio (from 0.5 to 0.55). The more efficient the converter, the smaller the needed duty ratio corrections for load current regulation. One motivation for the present invention is to separate the two disturbances by keeping the input voltage disturbance localized on the primary side of the isolation transformer T1 shown in FIG. 1 and load current disturbances localized on the secondary side of the isolation transformer T2 and thus eliminate the need for the isolation in the feedback control circuit.
Without such separation, the dynamic response and bandwidth of the original fourth order converter, such as the one in FIG. 5 is severely limited resulting in a poor transient performance to a step-load current change. In addition to the nonminimum phase response (right half plane zero's) the converter corner frequency is strongly dependent on the operating duty ratio D as discussed in S. Cuk and R. D. Middlebrook, Advances in Switched-Mode Power Conversion, Vols. I, II, and III, TESLAco, Irvine, Calif., 1981 and 1983 editions. Consequently, the loop-gain cross-over of the converter in FIG. 5 or other standard converters, such as the forward converter with an input filter shown in FIG. 4, has to be chosen at conservatively low frequencies in order to meet worst-case conditions.
Two approaches have been proposed in the past which provided a full-load regulation of multiple outputs puts in a single power conversion stage. The first approach described in A. Dauhajre and R. D. Middlebrook, A Simple PWM-FM Control for an Independently Regulated Dual Output Converter, Proc. Tenth International Solid-State Power Electronics, Conference (Powercon 10), March 1983, was based on a two-output flyback converter with one output operated in a discontinuous conduction mode, hence sensitive to switching frequency. The full regulation of both outputs was provided by controlling two quantities, duty ratio and the switching frequency of the single active device on the primary side. This method is clearly limited to two outputs and requires isolation in the feedback control circuit.
Another approach is to use additional active switches in each of the secondary converter sides of an isolation converter, such as the three-switch network (3SN) extension of the isolated Cuk converter proposed in R. Mahadevan, S. E1-Hamamsy, W.M. Polivka and S. Cuk, A CONVERTER WITH THREE SWITCHED-NETWORKS IMPROVES REGULATION, DYNAMICS, AND CONTROL, Proc. Tenth International Solid-State Power Electronics Conference (Powercon 10), pp. E1.1-E1.19, March 1983, and shown in FIG. 6. Note the additional pair of switches, Q1, D1 and Q2, D2 in each output section of the multiple output voltage regulator. As shown, each of the two outputs can be independently and fully controlled by independent PWM control signals q1 and q2. In waveforms shown in FIGS. 19 6a, which is equivalent to waveforms shown in FIG. 19 Mahadevan, et al., the arrows on the two drive waveforms for switches q1 and q2 indicate that these edges are controllable. Note the wide range of change of both duty ratios d1 and d2. Note also, that according to the control strategy proposed in Mahadevan, et al., supra duty cycle d of the PWM signal applied to the transistor Q on the primary side is constant, and its ability to vary is not utilized. Hence, both input voltage and load current variations are compensated by controlling the duty ratio of active devices only on the secondary winding side of the isolation transformer T. Thus, the same dynamic response deficiencies and sub-optimal step-load transient response remains. Furthermore, as illustrated in Mahadevan, et al., supra if the active device on the primary side is PWM modulated, it is controlled from the additional, standard, secondary-side output as shown in FIG. 28 in Mahadevan, et al., supra so that isolation in the feedback control is also needed.
A modification of the prior art in accordance with the present invention is to separate the conventional isolated feedback control loop of FIG. 5 into multiple, nonisolated feedback loops, one on the primary side of the isolated 3SN-Cuk converter of FIG. 6 for feedback regulation of the input voltage and the others for regulation of the multiple output voltages on its secondary side of the isolation transformer T. Thus, the beneficial decoupling of the input voltage regulation from the output load regulation is achieved, and each output voltage may be controlled by controlling the duty cycles d1 and d2 of the added third switches Q1 and Q2. The primary-side feedback loop regulates against input voltage variation, while the secondary-side feedback loops perform additional minor output voltage corrections against load current changes. This feedback decoupling is only possible on a very limited number of switching-mode converter topologies such as the 3SN-Cuk converter shown in FIG. 6.
In addition to elimination of the isolation in the feedback control, the invention to be described below provides full regulation of all output voltages in a single power processing stage while preserving the efficiency, small size, and low cost of the isolated Cuk converter disclosed in U.S. Pat. No. 4,184,197. Other switching-mode converters that may embody the present invention will then be described and illustrated in the drawings.
In ac-to-dc switching converters such as the one in FIG. 1, the conventional front-end, full-wave rectifier and filter provides raw dc input power which must be further processed by the dc-to-dc switching converter. The front-end, full-wave rectifier and filter has a very poor power factor and is a source of the severe harmonic currents. Recent international regulations governing power quality and harmonic current pollution of the utility line have imposed severe harmonic current limitations on equipment connected to the ac power line. Consequently, to avoid introducing harmonic currents in the line, it is desirable to operate a switching-mode converter with a unity power factor (UPF).
The standard way to achieve UPF operation is to process the input power as shown in FIG. 7 through a preliminary power processing stage 30, typically a boost converter which provides UPF control. The addition of this front-end preprocessing power stage further decreases efficiency and increases the size and cost of the converter package. It would be desirable to achieve UPF operation with only an appropriate modification of the input voltage regulation feedback loop in a single power processing stage. The present invention provides UPF operation in that manner without any penalty in efficiency, size, cost, or performance (fast output transient response), since UPF operation is obtained only through a change of the input feedback control loop of the single power processing stage of the converter which may be a multiple output voltage converter, each with its own direct feedback voltage control loop.